LED Current Balance Using a Variable Voltage Regulator with Low Dropout vDS Control

Abstract

A cost-effective light-emitting diode (LED) current balance strategy using a variable voltage regulator (VVR) with low dropout v_DS control is proposed. This can regulate the multiple metal-oxide-semiconductor field-effect transistors (MOSFETs) of the linear current regulators (LCR), maintaining low dropout v_DS on the flat v_GS-characteristic curves and making all drain currents almost the same. Simple group LCRs respectively loaded with a string LED are employed to implement the theme. The voltage VV_dc from a VVR is synthesized by a string LED voltage Nv_D, source voltage v_R, and a specified low dropout v_DS = V_Q. The VV_dc updates instantly, through the control loop of the master LCR, which means that all slave MOSFETs have almost the same biases on their flat v_GS-characteristic curves. This leads to all of the string LED currents being equal to each other, producing an almost even luminance. An experimental setup with microchip control is built to verify the estimations. Experimental results show that the luminance of all of the string LEDs are almost equal to one another, with a maximum deviation below 1% during a wide dimming range, while keeping all v_DS of the MOSFETs at a low dropout voltage, as expected.

1. Introduction

The improving of lighting efficiency is one of the direct ways to contribute to energy-saving and green environment initiatives. Furthermore, mercury-free requests for avoiding the release of pollution into the environment also form part of the vital work at present. Accordingly, light-emitting diodes (LEDs) have become the necessary option in the lighting environmental renovation. Even the power LED has advantages, including a high-fidelity, high rendering, and low power consumption. Currently, the most urgent task is to understand how to make LED luminance become a surface light source, such as a fluorescent lamp. The most effective way to do this is by using a much lower-power LED to implement the surface source. To date, effective control strategies for balancing LED currents only include the linear balancing way and the digital pulse balancing way, such as is shown in Figure 1 [1,2,3,4,5,6,7,8,9,10]; in which Figure 1a,b demonstrate linear balancing ways and Figure 1c illustrates the digital pulse balancing way. In Figure 1a, all LED strings are supplied by a constant voltage source (CVS). In this situation, the LED current balance can be easily determined by a voltage detected from a sense resistor. However, this technique may cause a sense voltage change, especially for the CVS supply, since the LED forward voltage will vary with the ambient temperature [11]. As shown in Figure 1b, a shunt current balance configuration is built with a constant current source (CCS) supply, in which all of the string LED currents are collected into the controller and then individually compared with the reference current. This can achieve more accurate current balancing between adjacent LEDs, in order to emit uniform light. In Figure 1c, a digital pulse shunt current balance circuit is implemented with a supply of either voltage source (VS) or current source (CS). In this case, LED current balancing mainly uses a PWM duty cycle control, such as sequential phase-shift duty control [12,13], burst mode [14,15], self-adaptive control [16], and series-connected mode [17], etc. In this case, the LED driven by the pulse to emit luminance for a human is based on the persistence of vision. Although the LED current balance by pulse drives can be conducted by multiplexing, it is difficult to make all of the string LED currents balanced for obtaining a uniform light source. To improve this problem, the luminance’s area should be fed back to adjust the adjacent string currents, producing a uniform light source. In spite of the fact that the mentioned current balance strategies are available, an uncertainty of the balance between the adjacent string LEDs always exists, mainly due to temperature variation and the dimming process. Further, the cost-effective way to perform a large scale lighting display is by using a shunt current balance configuration by linear current regulators (LCR) with a single voltage source [13,18]. Above all, the MOSFET of LCR may dissipate more conduction loss, due to an uncertain drain-to-source voltage on the i_DS-v_DS plane [19]. Besides, many balance cells for LEDs directly using versatile dc/dc converters or resonant converters have been successively developed [18,20,21,22,23,24,25]. However, in terms of application for a large scale display, it is difficult to implement a large lighting area using such a large amount LED cells, needed to acquire a balanced luminance. In fact, the cost-effective way to perform a large scale lighting display is to use multiple shunt current balance configurations, controlled by multiple linear current regulators (LCR) with only a single voltage source. In this paper, to improve the characteristics of LCRs for driving the string LEDs, all operation points of the MOSFETs are separately placed on their v_GS-controlled characteristic curves; curves which are produced from an array of IC values and which are expected to present correspondingly similar results [19]. On the specified almost-equal saturation-level curves, all string LEDs are then driven by almost constant drain currents to produce an equal luminance, even though the v_DS’s are scattered under the curves. Successively, an approach to place all MOSFET v_DS with a low dropout voltage on the specified flat curve in the saturation region is revealed [26]. Accordingly, a variable voltage regulator (VVR) is then proposed to perform the idea, such that all MOSFETs operate on the specified flat saturation curve with low dropout v_DS. The output voltage VV_dc of the VVR is synthesized by a string voltage Nv_D of N-LED, detected source voltage v_R, and a specified low dropout v_DS = V_Q. With the supply of VV_dc to LCRs, all string LEDs can be biased at almost the same saturation-level curves, respectively, whose currents will then be almost equal and can excite all string LEDs to produce an almost even luminance. Modeling the multiple LCRs for regulating constant string currents is conducted. An experimental setup for assessing the control strategies of balancing the multiple string LEDs is built. An aspect of the LCR for the string LED drives is described in Section 2. Using a variable voltage regulator (VVR) as the LCR supply, to excite the string LEDs for the production of a current balance, is explored in Section 3. The design and experiment for verifying the predicted estimations is discussed in Section 4. Final comments are concluded in Section 5.

2. Aspect of Linear Current Regulator for LEDs

2.1. Linear Current Regulator

Figure 2a shows the commonly-used LCR, which is a current-series feedback configuration composed of a MOSFET, an operational amplifier (OPA), and a sense resistor R_s. A string LED is widely loaded on the MOSFET in application. We define the symbols herein such that v_DS = V_DS + v_ds, where v_DS is for the total signal, V_DS is the dc component, and v_ds is the small signal. The small-signal model of the LCD is depicted in Figure 2b, in which the output current (drain current of MOSFET) i_DS is given by [17]. If A_v >> 1 and g_mA_vR_s >> 1, this yields:

$$i_{DS}\cong \frac{V_{+}}{R_s},$$

(1)

Presuming that the MOSFET operates in the saturation region, i_DS in (1) is almost a constant current, varying with the input voltage V₊. This drain current can also be directly found from the characteristic curve of the MOSFET, guided by a certain v_GS.

2.2. LED String Module in LCR

Bias Situation for a Single String LED

An LED is essentially a family of diodes, whose current is temperature-dependent and varies, approximately in exponential form, in its forward bias [19]. The relation of the general LED current i_D and the voltage v_D can be simply expressed by the Shockley equation; for a single LED, this is presented as:

$$i_D(v_D)=I_s(e^{\frac{v_D}{\eta V{}_T}}-1),$$

(2)

and

$$v(i_D)=v(i_{DS})=\eta V_T\ln (\frac{i_{DS}+I_s}{I_s}),$$

(3)

where I_s is the leakage current in reverse bias, V_T is the thermal voltage, η is the ideality factor, v_D is the forward voltage, and v_DS is the drain-to-source voltage of MOSFET. For a number of N LEDs connected in a series to form the string N-LED, the voltage v_DN of the forward bias can be given by:

$$\begin{array}{c}{v}_{DN}\equiv N{v}_D=\eta N{V}_T\ln (\frac{i_D+I_s}{I_s})\\ =\eta N{V}_T\ln (\frac{i_{DS}+I_s}{I_s})\end{array},$$

(4)

where i_DS = i_D is due to the series connection of N-LED, and N is an integer. With the assumption of the diode ideality factor η = 1 in standard fabrication, and if i_D >> I_s, we can simplify Equation (4) as:

$$\begin{array}{c}{v}_{DN}\approx N{V}_T\ln (\frac{i_D}{I_s})\\ \approx N{V}_T\ln (\frac{i_{DS}}{I_s})\end{array},$$

(5)

In Figure 2a, the loop equation for the LCR loaded with the string N-LED can be given by:

$$V_{dc}=v_{DS}+N{v}_D+i_{DS}{R}_s,$$

(6)

where V_dc is the dc supply voltage. The drain-to-source voltage of the MOSFET can be given by:

$$\begin{array}{c}{v}_{DS}=V_{dc}-N{v}_D-i_{DS}{R}_s\\ =V_{dc}-v\end{array},$$

(7)

where we let v be the voltage sum, i.e.,

$$v=N{v}_D+i_{DS}{R}_s,$$

(8)

Form (6), the drain current of the MOSFET in the LCR can also be given by

$$\begin{array}{c}{i}_{DS}=f\left(v_{DS}\right)\\ =\frac{1}{R_s}(V_{dc}-N{v}_D-v_{DS})\end{array},$$

(9)

For the convenience of analysis, point M is set to measure the drain voltage of the MOSFET, v_M, and Equation (6) can then be modified as:

$$\begin{array}{c}{v}_M=V_{dc}-N{v}_D\\ =v_{DS}+i_{DS}{R}_s\end{array},$$

(10)

If a small variation of v_D is neglected, V_M instead of v_M is given by:

$$\begin{array}{c}{V}_M=V_{dc}-N{V}_D\\ =V_{DS}+I_{DS}{R}_s\end{array},$$

(11)

Equation (11) represents a LCR loaded with one string N-LEDs and supplied by a voltage V_M = V_dc − NV_D. The combined characteristics of the one string N-LED and MOSFET of the LCR for bias and load-line descriptions, is shown in Figure 3a.

In this study, only the three kinds of temperature variations T₊ > T₀ > T₋ for the string N-LED are considered. Because of temperature-dependence, the voltage change of the N-LED will lead to a change in V_M, resulting in a move of the load line of MOSFET on the i_Ds-v_DS plane, which is clearly explored in Figure 3a. Correspondingly, the characteristics of N-LED on the i_D-v_D plane for the mentioned temperature variation are also depicted. In addition, the bias situation in the LCR with a constant V_dc supply is predictively shown in Figure 3b under temperature variation T₊ > T₀ > T₋, including the sense voltage v_Rs = i_DSR_s, Nv_D, v_DS, and the voltage sum v etc., all are presented with respect to the current i_DS = i_D.

(1) Ambient Temperature at T = T₀

In this situation, T = T₀, we presume that the MOSFET is initially biased at v_GS2. A thick-green dc load line of the MOSFET on the i_DS-v_DS plane is presented by (11), and the corresponding string N-LED characteristic curve on the i_D-v_D plane is given by (2), as respectively shown in Figure 3a. In other words, the operation point of MOSFET is at (I_DS20, V_DS20) and that of the N-LEDs is at (I_D20, V_D0), where I_DS20 = I_D20, while we presume that the point q₂₀ has a low dropout voltage V_DS20. For the load line on the i_DS-v_DS plane in this situation, the supply to the drain of the MOSFET is equivalent to V_M2 under the gate drive of v_GS2 in (11) and the V_DS2 = (V_M2 − V_Rs).

(2) Temperature falling to T = T₋

Once the environmental temperature T falls to T₋ from T₀, where T₋ < T₀, the string N-LED voltage will increase to V_D2 = Nv_D−, due to the negative temperature-dependent coefficient, and results in the V_M2− decreasing to V_dc − NV_D−. In this situation, the green-thick dc load line will move to the left, as a red line, leading the operation point q₂₀ moving left to q₂₋ while v_DS = V_DS2−, and while the MOSFET operates in the triode region and its drain current is i_DS = I₂₋; correspondingly, when the string N-LED current decreases, reducing its luminance, even its voltage V_D is slightly increasing. Accordingly, it may happen to push the operation point from the saturation region into the triode region, due to the increase of N-LED’s voltage in a lower temperature situation.

(3) Ambient Temperature rising to T = T₊

If the temperature increases to T₊ from T₀, where T₊ > T₀, the green-thick dc load line will move right, as a blue line, due to V_M2+ increasing to V_dc − NV_D+, which leads the operation point q₂₀ to move right to q₂₊ on the flat v_GS2− characteristic curve. Accordingly, the i_DS = I₂₊ ≈ I₂₀ and even the v_DS increases to V_DS2+; meanwhile, the string N-LED current is i_D = I₂₊ ≈ I₂₀, which keeps its luminance at almost the same value as that produced at I₂.

(4) Bias Situation for the MOSFET with Constant V_dc Supply for T₊ > T₀ > T₋

In order to explain the drain-to-source voltage of the MOSFET for T₊ > T₀ > T₋ under a constant voltage V_dc supply, a voltage sum v = Nv_D + i_DSR_s is introduced in Figure 3b. The voltage v, as the dark brown line, clearly displays the variation of v_DS between saturation and the triode regions, due to the temperature effect, where v_DS = V_dc − v.

2.3. Dimming with a Constant Voltage Supply to LCR

When using a LCR loaded with one string LED, with a constant supply of V_dc, for example, in the dimming process, the operation point of the MOSFET under different drives v_GS’s will change dramatically, since the voltage drop v_R on the source resistor R_s will vary far more than the voltage Nv_D. In this description, three kinds of gate voltages v_GS’s are explored, to dim the string N-LED. As mentioned previously, the operation of LCR is initially placed at point q₂ under normal temperature T₀, as shown in Figure 4, in which the N-LED current i_D = I_D2 under the MOSFET is driven by v_GS2. If the gate voltage changes from v_GS2 to v_GS1 while v_GS1 < v_GS2, the operation point q₂ will move to point q₁ on the flat v_GS1-characteristic curve, since the voltage Nv_D will decrease due to the decrease of i_D, lowering the LED luminous output. On the other hand, if the gate voltage varies from v_GS2 to v_GS3 for example, as well as v_GS3 > v_GS2, the voltage v_DS = V_DS3 will move left, to point q₃ on the triode region of the v_GS3-characteristic curve. In this instance, the drain current i_DS will increase to I_D3 since the voltage drop on R_s and Nv_D will increase, leading to the decrease of v_DS. Therefore, although an increase of the N-LED current i_D = I_D3 will raise the luminance, the luminance change may not be proportional to the dimming level, disturbing its uniform distribution variation.

3. Variable Voltage Regulator for LCRs to Balance the String LED Currents

3.1. VV_dc Synthesized to Clamp Low-Dropout v_DS in the MOSFET

As mentioned in Figure 3 and Figure 4, for a LCR with constant voltage V_dc supply, the drain-to-source voltage v_DS is not only dependent on the forward voltage of the string LED, but is also subject to the voltage drop v_R of the source resistor R_s. Accordingly, the v_R will increase, due to the increase of i_DS; the v_DS may fall dramatically to enter into the triode region of the MOSFET. On the contrary, the v_DS will be located far away from the triode region, due to the reducing i_DS, and the operation point will move right. Here, it will be sustained on the flat v_GS-characteristic curve, on which the drain current i_DS will remain almost constant in a wide variation of v_DS, resulting in a N-LED luminance output which is almost the same. In order to remedy the issue due to a constant voltage V_dc supply, a constant low-dropout v_DS is initially specified as V_Q, close to the boundary of triode region, but on the flat portion of the v_GS-characteristic curve. Subsequently, all drain voltages at points M, v_Mk, are only supplied by a variable dc supply VV_dck from a variable voltage regulator (VVR), in which the low dropout V_Q is involved, i.e.,

$$\begin{array}{c}{v}_{Mk}=V{V}_{dck}-N{v}_{Dk}\\ =V_Q+v_{Rk}\end{array},$$

(12)

where the footnote “k” denotes the kth operation point, and:

$$v_{Rk}=i_{DSk}{R}_s,$$

(13)

Then, the output voltage VV_dc of VVR can be estimated as:

$$\begin{array}{c}V{V}_{dck}=N{v}_{Dk}+V_Q+i_{DSk}{R}_s\\ =N{v}_{Dk}+v_{Mk}\end{array},$$

(14)

where the V_Q is a constant low voltage placed close to the boundary of the triode region, but in the saturation region, as shown in Figure 5a, i.e.,

$$\begin{array}{c}{V}_Q=V{V}_{dck}-v_k\\ \equiv \mathrm{constant}\end{array},$$

(15)

where the sum of the voltage v_k is given by:

$$v_k=N{v}_{Dk}+i_{DSk}{R}_s$$

(16)

Accordingly, the VV_dc can then be synthesized form (14), including Nv_D, i_DSR_s, and the specified low-dropout V_Q, as shown in Figure 5b. In Figure 5a, the supply v_Mk for different operation points under constant V_Q, can be easily found in (12). Remarkably, with a variable supply VV_dc, the LCR is able to keep the MOSFET operating at the specified V_Q under different v_GS’s, which are the corresponding dimming levels. In addition, with the low-dropout V_Q, the MOSFET can only dissipate low conduction loss, and the string N-LED can emit luminance, proportional to the dimming levels.

3.2. Dimming for Multiple-String N-LEDs

For dimming multiple-string N-LEDs, multiple identical i_DS-v_DS planes, as shown in Figure 6, are simultaneously controlled by multiple LCRs with varying v_GS values, in which all v_DS’s are then placed at a constant low-dropout v_DS = V_Q. For the convenience of analysis, all components, including MOSFETs, OPAs, LEDs, and source resistor R_s, are assumed to be identical in an array of integrated circuits (ICs), in which only one VV_dc, as shown in Figure 5b, is supplied in the planes of Figure 6. For the ease of analysis, two i_DS-v_DS planes are taken as being the same as each other, for example, in which V_Q1 and V_Q2 are the low-dropout voltages. Since the two MOSFETs are identical, the two operation points are almost the same, leading to the two string LEDs having the same currents. However, once there is a deviation from v_DS, at least one of the MOSFET’s v_DS should be maintained at the specified V_Q, leaving the rest of v_DS’s as being greater than V_Q, but still remaining on the same flat v_GS-characteristic curve; on which both of the drain currents of the MOSFETs in this example are almost the same, but are different for the v_DS. Thus, the two string N-LEDs can still have the same currents, and thus emit the same luminance. This control theme is to be extended to the current balance of multiple string N-LEDs.

3.3. Master and Slave Control for Current Balance of Multiple string N-LEDs

In this study, using an array of ICs to acquire identical MOSFETs for serving multiple-string N-LED control, is vital work. The circuit scheme for the multiple LCRs respectively loaded with one string N-LED, is outlined in Figure 7, in which the master and slave controls for the current balance are built. Since the devices and components, in practice, can be selected as almost identical to one another, the current balance between the adjacent string LEDs can then be realized as predictions. The control reference is preset at the master LCR and wires together with the other slave LCRs. The VV_dc of VVR is basically synthesized by the loop parameters of the master LCR, as per Figure 5b, including the v_DS of the MOSFET, v_R on source R_S, Nv_D on the N-LED, and the specified low voltage dropout V_Q. To implement the VVR, a reference voltage (V_Q + V_k)_ref is initially preset, where V_k depends on the state of the gate drive v_GS. After comparing the instantly detected voltage (v_k + v_DS) with the (V_Q + V_k)_ref in the master loop, the synthesized VV_dc is then adapted to supply all multiple string LEDs through the LCRs, and results in the same currents for adjacent LEDs, while holding all v_DS’s positions greater than the lowest V_Q. Once the control is finished in the master LCR, all slave LCRs with the updated supply VV_dc, can mirror the master drain current on their flat v_GS-characteristic curves in the i_DS-v_DS planes, and then all slave string LED currents are almost the same as one another.

3.4. Digitizing the VVR Controller

Figure 8 shows the digital control architecture used to build the VVR, in which the controller is executed by Microchip dsPIC33FJ06GS202, according to the idea theme in Figure 5b. The mentioned three parameters, v_DS, v_R, and Nv_D, detected from the master LCR are respectively sampled and held before the digitizing process. After combining them, a v_k + v_DS through D/A is obtained, to compare with the reference (V_Q + V_k)_ref at the error amplifier (EA). The error signal from the EA will then guide the VVR to provide an updated VV_dc for supplying the master and slave LCRs. This digitizing process makes the VV_dc produced from VVR more accurate and stable against the parameter variations mentioned previously.

4. Design and Experiment

To verify the proposed control theme, a scaled-down simple layout of 5 × 6 LEDs are arranged with five string LEDs connected in parallel, in which each string has six LEDs in series. Five LCRs are employed to drive the five string LEDs, respectively, in which MOSFET IRF110 is used as the current regulator in LCR, and the LED EHP-C04/UT01-P01/TR with a rated power of 1 W is adopted. The operational amplifier TL074 with a slew rate of 13 V/μs is used as a comparator. The power LED has a conductance (i_D/v_D) of approximately 0.42 mA/mV. The transconductance (i_DS/v_GS) of the MOSFET is about 2.97 mA/mV under an i_DS varying from 0 mA to 350 mA; correspondingly, v_GS varies from 3.72 V to 4.5 V. The low-dropout v_DS = V_Q, is 1.5 V, specified at i_DS ≈ 200 mA. The circuit schemes for the experiment are as in Figure 7 and Figure 8, in which the digital controller for synthesizing the VVR is as per the loop parameters of the master LCR, with reference to the estimation profile in Figure 5b. In Figure 7, the v_DS is instantly detected by the differential amplifier (A1) and is controlled at a low dropout V_Q. The voltage v_k = f(i_DSk) in (16) is a function of the kth current i_DSk detected from a source voltage (v_R), in which the voltage v_k is the summation of v_R and Nv_D, where the footnote “k” is used to deonte the position of the operation point driven by v_GSk. The v_Q and v_k are merged at the inverting input of the error amplifier (A2) and compared with a specified reference voltage (V_Q + V_k)_ref. After comparing the error output V_ea from A2 with a reference sawtooth at the PWM controller (A3), an updated variable voltage VV_dc from the VVR is acquired through the loop regulation in the master LCR. With the VV_dc, the v_DS’s in the five string LCRs can then be kept at a low dropout V_Q on the five flat v_GS-characteristic curves, respectively. A source resistor R_s = 12 Ω which is, estimated for the current sensor is used, the VVR is supplied by a dc source V_dc = 35 V, and the dimmer control is implemented by a precision variable resistor. The digitizing controller, as shown in Figure 8, is implemented by Microchip dsPIC33FJ06GS202, in which three parameters, including v_D5, v_R, and v_DS, detected from the control loop of the master LCR, are sampled and held for digitizing. Through D/A conversion, the voltage v_k is acquired. After merging the v_k and v_DS to compare with the reference (V_Q + V_k)_ref at the error amplifier (EA), the updated VV_dc from the VVR to supply the five LED strings, is achieved. Consequently, the self-feedback in the control loop of the master LCR for keeping all slave v_DS’s at low dropout is always valid, and indeed the updated VV_dc is able to achieve the current balance in the multiple LED strings.

4.1. Experiment for a Single String 6-LED

The experiment to evaluate the characteristics of a LCR to drive a single string 6-LED and their relative parameters during v_GS changes, is shown in Figure 9. The synthesis voltage VV_dc composed of v_DN = Nv_D, V_R, and v_DS, under the v_GS control, is displayed in Figure 9a, in which all of the parameters measured meet the estimations with an acceptably low dropout voltage. The experiment shows that v_DS varies at around 1.5 V, from 1.77 V to 1.38 V, in all five MOSFETs; correspondingly, their v_GS varies from 3.72 V to 4.46 V.

The LED luminance, with respect to the LED current i_D under the v_GS control, is shown in Figure 9b, in which each string 6-LED can emit a luminance from 0 to 62.1 kcd/m²; correspondingly, the string LED current i_D varies from 0 to 357 mA. The string LED luminance is measured vertically, facing the LED display at a distance of 50 cm, by using a luminance-meter type BM-910 TOPCON with a high accuracy. The string LED luminance and the conduction loss in MOSFET are measured in Figure 9c. It is clearly seen that the maximum conduction loss in the MOSFET is only 492 mW at i_DS = 357 mA, due to a low dropout of v_DS ≈ 1.38 V, which actually benefits the MOSFET, dissipating the low conduction loss.

4.2. Experiment for the Multiple String 6-LEDs

In this experiment, the master and the four slave LCRs are supplied by the same VV_dc, and all of the controls are wired with the same dimmer signal. The VV_dc is controlled by the loop of the master LCR. With the VV_dc, each slave LCR can self-regulate the MOSFET’s v_DS to stay at the low dropout V_Q, even at place higher than V_Q toward right-side, where the drain current can always mirror the master current during the dimmer control.

The variations of the five v_DS’s, with respect to each string’s LED current, are shown in Figure 10a, in which the highest v_DS is 1.78 V and the lowest v_DS is 1.37 V, and all are located on the flat v_GS-characteristic curves. With reference to v_DS = V_Q = 1.5 V, the deviation ∆v_DS above V_Q is 0.28 V and below V_Q is 0.13 V, in which the low deviation is still on the flat v_GS-characteristic curve in the saturation region of MOSFET. The luminance’s distribution of the 5-group LED strings during the LED current change is displayed in Figure 10b, in which the adjacent luminance values between 6-LED strings are almost the same during a wide dimmer range. In this measurement, the maximum luminance deviation, of about ±0.45 kcd/m², occurs at the highest luminance of 62.7 kcd/m² (where i_DS = 357 mA) and the minimum deviation of about ±0.01 kcd/m² at the lowest 2.65 kcd/m² (where i_DS = 10 mA). The measured luminance can be expected to be uniform if the proposed 5 × 6 LED layout is equipped in a panel display. This experiment successfully demonstrates the current balance strategy for the multiple string LED; particularly, the control theme, which is easy to implement by using simple LCRs with a microchip control. The experimental setup of a scale-down 5 × 6 LED display is shown in Figure 11.

5. Conclusions

In this paper, the synthesized VVR successfully achieves a low dropout v_DS for all MOSFETs in the LCRs, and allows their operation points to remain on the flat v_GS-characteristic curves, on which all of the drain currents are almost the same and actually attain the balance of the multiple string LED currents. In addition, the experiment evidenced that the control strategy is feasible in practice. All measurements, including the current balance behavior, low dropout voltage deviation, and the distribution of luminance between the adjacent string LEDs etc., all meet the estimations. Consequently, the cost-effective LCRs with the proposed control theme can actually achieve the current balance in multiple string LEDs, and it would be suitable to extend their use to a large-scale display that requires multiple string LEDs.